1. Field of the Invention
The invention relates to a direct conversion receiver, and in particular, to a calibration method for improving IIP2 characteristics in the direct conversion receiver.
2. Description of the Related Art
FIG. 1 shows a conventional 2nd order input intercept point (IIP2) characterization. For a down-converter in a direct conversion receiver, RF input power and IF output power are measured. Input power is plotted along the horizontal axis, and output power is plotted along the vertical axis. The most dominant 2nd order distortion source in a receiver is the down conversion stage, and a 2nd order input intercept point (IIP2) is defined as a performance index. Mixers used in integrated direct conversion receivers are usually active transconductance mixers and nearly always balanced or double-balanced structures. A high IIP2 point is desirable for a direct conversion receiver. In a perfect balanced case, the IIP2 is infinite. Practically, however, mismatches of the components are inevitable, reducing the IIP2. Thus, compensation for the mismatches is required.
FIG. 2 shows a conventional mixer. A typical mixer model utilizes a differential loading pair comprising a 1st load and a 2nd load. Conventionally, mismatches of the components are unavoidable. For example, duty cycles of the local oscillation signals VLO+ and VLO−, amplitudes of the RF signals VRF+ and VRF−, parameters of 1st switch and 2nd switch, and resistances of 1st load and 2nd load may comprise erroneous inaccuracy, represented as follows:
                                                        A                              RF                +                                      =                                          A                RF                            ⁡                              (                                  1                  +                                                            Δ                      ⁢                                                                                          ⁢                                              A                        RF                                                              2                                                  )                                              ;                ⁢                                  ⁢                              A                          RF              -                                =                                    A              RF                        ⁡                          (                              1                -                                                      Δ                    ⁢                                                                                  ⁢                                          A                      RF                                                        2                                            )                                                          (        1        )            
where ARF+ and ARF− are amplitudes of the RF signals VRF+ and VRF−, and ΔARF is their difference.
                                                        g                              m                +                                      =                                                            g                  m                                ⁡                                  (                                      1                    +                                          Δ                      ⁢                                                                                          ⁢                                              g                        m                                                                              )                                            2                                ;                ⁢                                  ⁢                              g                          m              -                                =                                                    g                m                            ⁡                              (                                  1                  -                                      Δ                    ⁢                                                                                  ⁢                                          g                      m                                                                      )                                      2                                              (        2        )            
where gm+ and gm− are conductivities of the components in 220, and Δgm is their difference.
                                                        η              +                        =                                          η                nom                            ⁡                              (                                  1                  +                                      Δη                    2                                                  )                                              ;                ⁢                                  ⁢                                            η              -                        =                                          η                nom                            ⁡                              (                                  1                  -                                                            Δ                      ⁢                                                                                          ⁢                      η                                        2                                                  )                                              ;                ⁢                                  ⁢                              η            nom                    =                      50            ⁢            %                                              (        3        )            
where η+ and η− are duty cycles of the local oscillation signals VLO+ and VLO−, and Δη is their difference.
                                                        R                              L                +                                      =                                          R                L                            ⁡                              (                                  1                  +                                                            Δ                      ⁢                                                                                          ⁢                      R                                        2                                                  )                                              ;                ⁢                                  ⁢                              R                          L              -                                =                                    R              L                        ⁡                          (                              1                -                                                      Δ                    ⁢                                                                                  ⁢                    R                                    2                                            )                                                          (        4        )            
where RL+ and RL− are the 1st load and 2nd load, and ΔR is their difference.
These mismatches are factors causing IIP2 reduction. Various calibration methods can compensate the mismatches. In an IEEE paper “Characterization of IIP2 and DC-Offsets in Transconductance Mixers”, IIP2 is calculated as functions of load resistor imbalance and duty cycle mismatch, and the ΔR is tuned to optimize the IIP2 of a mixer. In another IEEE paper, Young-Jin Kim, “A GSM/EGSM/DCS/PCS Direct Conversion Receiver With Integrated Synthesizer”, an adjustable resistor is provided for coarse and fine calibrations of the load mismatch. The load mismatch varies as a digital code of 8 bits. The variation of the load mismatch, however, is not linear to the digital code values, thus a wide range of trial digital codes are required to locate an optimum result. Further, the nonlinearity of the adjustable resistor may not permit sufficient accuracy for the mismatch compensation. Therefore, an enhanced architecture and calibration method are called for.